The present invention relates generally to high impedance surfaces. More particularly, the present invention relates to artificial magnetic conductor surfaces loaded with ferrite-based artificial magnetic materials.
A high impedance surface is a lossless, reactive surface whose equivalent surface impedance,
            Z      s        =                  E        tan                    H        tan              ,approximates an open circuit and which inhibits the flow of equivalent tangential electric surface current, thereby approximating a zero tangential magnetic field, Htan≈0. Etan and Htan are the electric and magnetic fields, respectively, tangential to the surface. High impedance surfaces have been used in various antenna applications. These applications range from corrugated horns which are specially designed to offer equal electric (E) and magnetic (H) plane half power beamwidths to traveling wave antennas in planar or cylindrical form. However, in these applications, the corrugations or troughs are made of metal where the depth of the corrugations is one quarter of a free space wavelength, λ/4, where λ is the wavelength at the frequency of interest. At high microwave frequencies, λ/4 is a small dimension, but at ultra-high frequencies (UHF, 300 MHz to 1 GHz), or even at low microwave frequencies (1-3 GHz), λ/4 can be quite large. For antenna applications in these frequency ranges, an electrically-thin (λ/100 to λ/50 thick) and physically thin high impedance surface is desired.
One example of a thin high-impedance surface is disclosed in D. Sievenpiper, “High-impedance electromagnetic surfaces,” Ph.D. dissertation, UCLA electrical engineering department, filed January 1999, and in PCT Patent Application number PCT/US99/06884. FIG. 1 shows an example of such a high impedance surface 100. The high-impedance surface 100 includes a low permittivity spacer layer 104 and a capacitive frequency selective surface (FSS) 102 formed on a metal backplane 106. Metal vias 108 extend through the spacer layer 104, and connect the metal backplane to the metal patches of the FSS layer, creating what may be termed a thumbtack structure. The thickness h of the high impedance surface 100 is much less than λ/4 at resonance, and typically on the order of λ/50, as indicated in FIG. 1.
The FSS 102 of the prior art high impedance surface 100 is a periodic array of metal patches 110 which are edge coupled to form an effective sheet capacitance. This is referred to as a capacitive frequency selective surface (FSS). Each metal patch 110 defines a unit cell which extends through the thickness of the high impedance surface 100. Each patch 110 is connected to the metal backplane 106, which forms a ground plane, by means of a metal via 108, which can be plated-through holes. The periodic array of metal vias 108 has been known in the prior art as a rodded media, so these vias are sometimes referred to as rods or posts. The spacer layer 104 through which the vias 108 pass is a relatively low permittivity dielectric typical of many printed circuit board substrates. The spacer layer 104 is the region occupied by the vias 108 and the low permittivity dielectric. The spacer layer is typically 10 to 100 times thicker than the FSS layer 102. Also, the dimensions of a unit cell in the prior art high-impedance surface are much smaller than λ at the fundamental resonance. The period is typically between λ/40 and λ/12. This configuration of metal patches 110 and metal vias 108 may be referred to as a thumbtack structure.
A frequency selective surface (FSS) is a two-dimensional array of periodically arranged elements which may be etched on, or embedded within, one or multiple layers of dielectric laminates. Such elements may be either conductive dipoles, patches, loops, or even slots. As a thin periodic structure, an FSS is often referred to as a periodic surface.
Frequency selective surfaces have historically found applications in out-of-band radar cross section reduction for antennas on military airborne and naval platforms. Frequency selective surfaces are also used as dichroic subreflectors in dual-band Cassegrain reflector antenna systems. In this application, the subreflector is transparent at frequency band f1 and opaque or reflective at frequency band f2. This allows placement of a feed horn for band f1 at the focal point for the main reflector, and another feed horn operating at f2 at the Cassegrain focal point. In this manner, a significant weight and volume savings can be achieved over using two conventional reflector antennas. Such savings is critical for space-based platforms.
The prior art high-impedance surface 100 provides many advantages over corrugated metal structures. The surface is constructed with relatively inexpensive printed circuit technology and can be made much lighter than a corrugated metal waveguide, which is typically machined from a block of aluminum. In printed circuit form, the prior art high-impedance surface can be 10 to 100 times less expensive for the same frequency of operation. Furthermore, the prior art surface offers a high surface impedance for both x and y components of tangential electric field, which is not possible with a corrugated waveguide. Corrugated waveguides offer high surface impedance for one polarization of electric field only. According to the coordinate convention used herein, a surface lies in the x-y plane and the z-axis is normal or perpendicular to the surface. Further, the prior art high-impedance surface provides a substantial advantage in its height reduction over a corrugated metal waveguide, and may be less than one-tenth the thickness of an air-filled corrugated metal waveguide.
A high-impedance surface is important because it offers a boundary condition which permits wire antennas conducting electric currents to be well-matched and to radiate efficiently when the wires are placed in very close proximity to this surface (e.g., less than λ/100 away). The opposite is true if the same wire antenna is placed very close to a metal or perfect electric conductor (PEC) surface. The wire antenna/PEC surface combination will not radiate efficiently due to a very severe impedance mismatch. The radiation pattern from the antenna on a high-impedance surface is confined to the upper half space, and the performance is unaffected even if the high-impedance surface is placed on top of another metal surface. Accordingly, an electrically-thin, efficient antenna is very appealing for countless wireless devices and skin-embedded antenna applications.
Another example of a high impedance surface is disclosed in U.S. Pat. No. 6,512,494 B1, issued to Diaz, et al. on Jan. 28, 2003. This reference discloses an artificial magnetic conductor which is resonant at multiple resonance frequencies. The artificial magnetic conductor is characterized by an effective media model which includes a first layer and a second layer. Each layer has a layer tensor permittivity and a layer tensor permeability having non-zero elements on the main tensor diagonal only. U.S. Pat. No. 6,512,494 B1 is incorporated herein in its entirety by this reference. The disclosed AMC is a two-layer, periodic, magnetodielectric structure where each layer is engineered to have a specific tensor permittivity and permeability behavior with frequency. This structure has the properties of an artificial magnetic conductor over a limited frequency band or bands, whereby, near its resonant frequency, the reflection amplitude is near unity and the reflection phase at the surface lies between +/−90 degrees. This engineered material also offers suppression of transverse electric (TE) and transverse magnetic (TM) mode surface waves over a band of frequencies near where it operates as a high impedance surface.
FIG. 2 is a photograph of a prior art artificial magnetic conductor 200. The AMC 200 is embodied with a thick foam core spacer layer 204 and an array of metal patches 210 with metal vias 208 extending from some of the metal patches 210 through the spacer layer 204. The AMC 200 was developed under DARPA Contract Number F19628-99-C-0080. The size of the AMC 200 is 10 in. by 16. in by 1.26 in thick (25.4 cm×40.64 cm×3.20 cm). The weight of the AMC is 3 lbs., 2 oz. The 1.20 inch (3.05 cm) thick, low permittivity spacer layer is realized using foam. The FSS has a period of 298 mils (7.57 mm), and a sheet capacitance of 0.53 pF/sq.
FIG. 3 shows the measured reflection coefficient phase referenced to the top surface of the AMC 200 as a function of frequency. A ±90° phase bandwidth of 900 MHz to 1550 MHz is observed. Three curves are traced on the graph, each representing a different density of vias within the spacer layer (one out of every two possible vias is installed, curve AMC 1-2, one out of every four is installed, curve AMC 1-4, and one out of every 18 vias is installed, curve AMC 1-18). As expected from the effective media model described in U.S. Pat. No. 6,512,494 B1, the density of vias does not have a strong effect on the reflection coefficient phase.
Test set-ups are used to experimentally verify the existence of a surface wave bandgap in an AMC. In each case, the transmission response (S21) is measured between two Vivaldi-notch radiators that are mounted so as to excite the dominant electric field polarization for TE and TM modes on the AMC surface. For the TE set-up, the antennas are oriented horizontally. For the TM set-up, the antennas are oriented vertically. Absorber is placed around the surface-under-test to minimize the space wave coupling between the antennas. The optimal configuration—defined empirically as “that which gives us the smoothest, least-noisy response and cleanest surface wave cutoff”—is obtained by trial and error. The optimal configuration is obtained by varying the location of the antennas, the placement of the absorber, the height of absorber above the surface-under-test, the thickness of absorber, and by placing a conducting foil wall between layers of absorber.
FIG. 4 illustrates the measured S21 for both transverse electric (TE) and transverse magnetic (TM) configurations for the AMC 200 of FIG. 2. As can be seen, a sharp TM mode cutoff occurs near 950 MHz, and a gradual TE mode onset occurs near 1550 MHz. This bandgap is correlated closely to the +/−90-degree reflection phase bandwidth of the AMC.
Broadband antennas such as spirals can be mounted over the thick foam core AMC 200 of FIG. 2. FIG. 5 shows a spiral antenna on the thick foam AMC core of FIG. 2. Such antennas exhibit good impedance and gain performance over the range of frequencies where both a +/−90-degree reflection phase occurs, for normal incidence, as well as where a surface wave bandgap (where both TM and TE modes are cutoff) is found.
In most wireless communications applications, it is desirable to make the antenna ground plane as small and light weight as possible so that it may be readily integrated into physically small, light weight platforms. The relationship between the instantaneous bandwidth of an AMC such as the AMC 200 of FIG. 2 and its thickness is given by the following equation:
      BW          f      0        =      2    ⁢          πμ      r        ⁢                  ⁢                  h                  λ          0                    .      
Here, h is the thickness of the spacer layer, λ0 is the free space wavelength at resonance where a zero degree reflection is observed and μr is the magnetic permeability of the spacer layer. As can be seen from this equation, to support a wide instantaneous bandwidth BW. the AMC thickness λ0 must be relatively large or the permeability must be high μr. For example, to accommodate an octave frequency range (BW/f0=0.667), the AMC thickness must be at least 0.106 λ0, corresponding to a physical thickness of 1.4 inches (3.56 cm) at a center frequency of 900 MHz. This thickness is too large for many practical applications. As noted, the antenna ground plane should be as small and light weight as possible.
Accordingly, there is a need for an improved artificial magnetic conductor with enhanced bandwidth offering reduced size and weight.